1. Field of the Invention
This invention relates to coherent ultrasound imaging systems and, more particularly, to phased array ultrasound imaging systems operating in different scan formats and imaging modes. Specifically, but not limited to, the invention relates to phased array beamformer system with low-noise Doppler data acquisition.
2. Description of the Related Art
Medical ultrasound imaging systems are capable of many different modes of operation. One of these is the Doppler mode dedicated to displaying the movement of blood within a vein or an artery.
Doppler imaging can be performed using either continuous wave (CW) or pulse wave (PW) techniques. In CW Doppler acquisition, the ultrasound transmitter continuously insonifies the body, while the receiver continuously receives echoes from all objects within the receiver's area of sensitivity. In this case, information received from any specific range interval cannot be isolated. Accordingly, the observation region is the overlap portion between the transmitting and receiving transducer beam profiles. To select the desired target, the instrument's area of sensitivity is adjusted, by either physical placement of the probe, by beamforming, or both.
As a single scatterer passes the observation region, the scatterer generates a burst of oscillations that contributes to the received radio frequency (RF) signal. The frequency of this oscillation is different from the transmit frequency because of the Doppler shift, which is proportional to the component of the blood velocity along the phase gradient of the combined transmitter and receiver beams. The “sign,” or relative polarity, of the frequency difference between the transmitted and received signals determines the direction of the blood flow.
In PW mode, the scanner transmits a periodic pulse wave at a certain operating frequency F0 that is directed to a particular location having blood flow. The signal reflecting from the moving blood is shifted in frequency by an amount proportional to the velocity of the blood flow. Thus, with the PW Doppler technique, the received signal has the same essential properties as for the CW. The difference is that the range gate limits the observation region along the beam to the range cell. This allows one to obtain only samples of the Doppler signal with the pulse repetition frequency, the PRF, which introduces the problem of frequency aliasing. Besides, due to limitations in the PRF rate, PW Doppler has limited ability to measure very high blood flow velocities. The rate limitations are fundamental because the transmitted pulse must reach the target and echoes are reflected back to the receiver before the next pulse can be sent.
CW Doppler, on the other hand, transmits a constant continuous wave signal toward the area to be imaged at a particular transducer operating frequency. The signal is continuously reflected by the blood flow and received by a receiver. The receiver distinguishes between the transmitted signal and the received signal by determining if there is a frequency shift between the transmitted and received signals. The movement of the blood causes this frequency shift, where its value is proportional to the velocity of the blood. The direction of the blood flow is dependent on whether the frequency of the received signal is greater or less than the frequency of the transmitted signal. Because the signal is transmitted continuously, CW Doppler can detect significantly higher frequency shifts than PW Doppler since there is no inherent sampling rate limitation.
There are two sources contributed to an RF signal received from an internal structure of human body containing a blood vessel. First, this is a strong signal from slowly moving tissue with low Doppler shifts (0 Hertz for stationary tissue). The Doppler signals from blood can be 60-100 dB weaker exhibiting larger shifts in frequency because the blood has higher velocities than the tissue. FIG. 2a illustrates these differences.
In comparison with PW Doppler, it is more difficult for CW techniques to distinguish between the transmitted signal and the reflected signal originated by moving blood. First, since the transmitted signal is continuous and relatively high in amplitude, it generates interference in the receiver. Second, the high-amplitude echoes reflected from stationary tissue, typically called “clutter,” do not contain a frequency shift but occur simultaneously with the signals that represent blood flow. Further, conventional CW Doppler processors have a limited dynamic range due to the limited dynamic range of the analog-to-digital converters (ADCs). Thus, the clutter filtering that precedes the ADC must be much more complex, so that the signal that feeds the converter contains as little clutter content as possible.
There are numerous methods and techniques that have been developed to enhance quality of Doppler data acquisition. The following U.S. Patents represent typical examples of prior art, merely by way of example: U.S. Pat. Nos. 4,866,613, 4,911,171, 5,555,534, 5,562,097, 6,544,180, 6,527,722, and 6,648,826.
In general, the separate analog-processing path for a CW Doppler receiver consists of cascaded stages of mixers and filters. To support a variety of transducers, the hardware includes a number of programmable filters that are tuned to the operating frequencies of the available transducers. Such architecture requires using expensive switches and precision components. For the phased array, i.e., multi-channel ultrasound systems, it causes a substantial increase in the component count and cost that makes this approach impractical.
To address the complexity issue, Fazioly, et al., U.S. Pat. No. 6,527,722, describes a CW Doppler single channel receiver consisting of a mixer accompanied by a bandpass filter (BPF), which operates to translate the RF input signal to a constant intermediate frequency (IF) signal. Consequently, the cost of the CW Doppler processing circuitry will be reduced with respect to a conventional processing system. However, referring to a single-channel receiver, Fazioly does not disclose any aspect of CW Doppler data acquisition with a phased-array transducer.
Maslak, et al., U.S. Pat. No. 5,555,534, teaches a phased array receive beamformer that is dedicated to operate in both CW and PW Doppler modes. FIG. 1 depicts a block diagram of the beamformer comprising a plurality of receive channels 110. Each of the channels includes a low-noise amplifier (LNA), a gated quadrature mixer, and a complex rotator. In operation, the RF signal amplified by LNA 111 is mixed in a quadrature mixer with a pair of clocks being out of phase by 90° with respect to each other. The in-phase clock signal LOI, which is supplied to mixer 102, is provided in common to the in-phase mixers of all of the analog receive channels 110, as is the quadrature clock signal LOQ received by mixer 104. The outputs of the mixers 102 and 104 are in-phase and quadrature-phase components of a complex baseband signal related to respective RF echo. These outputs are coupled to a complex rotator 106, which is a baseband signal processing block, that weights, selects, and sums the in-phase and quadrature-phase components. The I/Q outputs of the rotator are programmed to represent eight possible phases of the input complex signal. The rotator in each channel has its own set of three phase control input bits.
Referring to FIG. 1 again, the in-phase (I) signals 108 of all of the individual Doppler receive beamformer channels 110 are summed in four groups. At first, the per-group signals 108 are applied to respective summers 112 having a low-pass pole 114, which filters out the RF products of the mixing process without affecting the baseband component. Then, the partial sums 118 are combined by a summer 116 to generate a beamformed in-phase signal 120 from all channels. It will be understood that the quadrature signals are combined in the same manner.
The outputs of the I/Q summers are coupled to a downstream processor 140. The processor comprises in-phase and quadrature sections but since they are identical, only the in-phase section is shown. It includes an integrator 122 to integrate (PW) or to smooth (CW) the beamformed signals, a track-and-hold circuit 124, a high-pass filter 126 to remove clutter signals, an anti-aliasing filter 128, and an ADC 130 to convert the relatively clutter-free signals to digital format.
As known in the art, there are indisputable advantages of the baseband representation of composite RF signals similar to those shown in FIG. 2a. However, since blood flow originates a Doppler shift in the audible range, the spectrum of the baseband I/Q components occupies the same frequencies as flicker or 1/f noise. For reasons, which will become apparent, the overlapping of spectra as shown in FIG. 2b may substantially reduce the dynamic range of D-mode acquisition.
There are two main sources of the 1/f noise in the '534 beamformer:
First, the per-cannel quadrature mixers, 102 and 104, comprise two transistor pairs switching at the LO frequency. While switching, the gates (bases) of the pair exhibits charge fluctuations. Having a spectral density proportional to 1/f, these fluctuations are transferred to the output by multiplication with a time-varying transconductance of the switching pair. Since transconductance of the pair is varied at the 2×LO frequency, it contains only even-order harmonics of the LO. This means that flicker noise from the switching pair will directly appear at the output around DC, i.e., in baseband. (Sometimes this noise is referred as the phase noise.) For an N-channel beamformer, the resulting 1/f noise from switching is increased for a factor of N1/2 as compared with a single channel. However, since the beamformer signal gain is equal to N, the signal-to-noise ratio (SNR) is improved by a factor of N1/2.
Another source of flicker noise is subsequent summing inherently associated with the process of beamforming. Referring to '534 in particular, the complex rotator 106 sums the weighted baseband outputs of the mixers 102 and 104. It is followed by the combining of all of the per-channel I/Q output signals represented in the baseband. The noise-referred details of the summing operation are discussed below.
It will be evident to those skilled in the art that the LNA/mixer combination needs to provide a gain, which is sufficient to prevent substantial degradation of the SNR by the noise introduced by subsequent summing means. However, a weak signal representing blood flow is situated on a top of a high-amplitude clutter, which may be in the range of 500 millivolts peak-to-peak. Consequently, the entire signal-processing chain needs to be relatively high-voltage in order to avoid signal clipping.
By contrast, the latest integration technology is based on low-voltage MOS processes with signal swing of 1.8 Volts or less. Thus, developing an integrated MOS receiver for D-mode, the LNA/mixer gain may not exceed 12 dB. For a given gain G, the expected SNR degradation due to 1/f noise can be found as follows:
Let fC denote flicker noise corner frequency, i.e., the frequency at which 1/f noise exceeds thermal noise. Depending on the operating conditions of the fabrication process, MOS devices manifest a corner frequency, which varies as the reciprocal of the channel length. Typically, fC has a range of 100 kHz to 1 MHz.
If SF and ST are the power spectral densities of 1/f noise and thermal noise, respectively, their densities can be equated at a corner frequency of fC, i.e., SF (fC)=ST. By definition, the power spectral density of 1/f noise is SF=K/f. Resolving this formula for f=fC yields K=ST·fC. It allows to express flicker noise spectrum as SF=ST·fC/f.
In addition to the phase noise, noise contribution from the LNA/mixer section is primarily related to translating LNA noise from the RF range (2 to 8 MHz, typically) to the baseband. Since the above RF range is well above the 1/f noise corner, the corresponding noise is representative of thermal noise. This noise manifests a noise floor for a subsequent stage, i.e., the summer 112. The noise-floor spectrum introduces by the LNA/mixer section is relatively flat with power spectral density of G2·ST. Because noise contributions from the switching pair and said noise floor are mutually independent, their influence can be considered separately.
At the summing node of 112, the resulting noise exhibits a linear combination of the above-mentioned noise floor and the input referred noise produced by the summer itself. If we consider the input referred noise to be originated by thermal and flicker sources, the total noise power can be expressed as the sum of three definite integrals, each related to respective noise source. To determine the limits of integration, it can be taken into account that the clutter filter removes any Doppler along with noise components occurring at or near 0 Hertz. Let fMIN denote a minimal frequency of a signal passing the clutter filter. On the other end, the highest Doppler shift determines a cut-off frequency of the processing, fMAX. Typically, fMAX=100 kHz. Thus, the total noise power, VN2, yields:
      V    N    2    =                    ∫                  f          MIN                          f          MAX                    ⁢                        S          T                ⁢                                  ⁢                  ⅆ          f                      +                  G        2            ×                        ∫                      f            MIN                                f            MAX                          ⁢                              S            T                    ⁢                                          ⁢                      ⅆ                          +                                                ∫                                      f                    MIN                                                        f                    MAX                                                  ⁢                                                      S                    F                                    ⁢                                                                          ⁢                                      ⅆ                    f                                                                                          Evaluating the integrals,
      V    N    2    =            S      T        ⁡          [                                    (                          1              +                              G                2                                      )                    ·                      (                                          f                MAX                            -                              f                MIN                                      )                          +                                            f              C                        ·            ln                    ⁢                                          ⁢                                    f              MAX                                      f              MIN                                          ]      In the absence of flicker noise, the total noise power would be VNT2, where:VNT2=ST·(1+G2)·(fMAX−fMIN)Taking the ratio of VN to VNT, the SNR degradation due to 1/f noise of the summer can be expressed as:
  γ  =                    V        N        2                    V        NT        2              =          1      +                                                  f              C                                                      (                                                      f                    MAX                                    -                                      f                    MIN                                                  )                            ⁢                              (                                  1                  +                                      G                    2                                                  )                                              ·          ln                ⁢                                  ⁢                              f            MAX                                f            MIN                              
TABLE I sets forth the amount of SNR degradation, γ, for fMIN=1, 10, 100, and 1000 Hz with G=4 and fC=1 MHz.
TABLE Ifmin (Hz)1101001000γ(dB)8.9068.0747.0445.692It can be seen that flicker noise associated with subsequent summing stages increases the system noise floor by a factor of 7-8 dB that substantially degrades the performance of beamforming provided in the baseband.
As mentioned, the influence of 1/f noise can be minimized by increasing the LNA/mixer gain. However, in a low-voltage system this approach is practically inapplicable since clutter would desensitize the beamformer before the signals from blood flow become sufficiently large. Therefore, there is a need for a wide-dynamic-range phased array Doppler beamformer adapted to operate using sub-micron technology.